A voltage rectifier acts as a switch that has a low resistance to current flow in a first voltage/current quadrant, and a very high resistance to current flow in the three remaining voltage/current quadrants. Referring to FIG. 1a, characteristics of an ideal voltage rectifier are illustrated. An ideal voltage rectifier acts as a switch that imposes zero resistance to current flow in the first voltage/current quadrant, and imposes infinite resistance to current flow in quadrants two through four. As illustrated in FIG. 1b, a practically achievable passive semiconductor diode exhibits less than ideal characteristics.
An active circuit employing a feedback amplifier and power switching device such as a MOSFET transistor may be used instead of a passive diode so as to realize a transfer function that is a much closer approximation of an ideal diode in quadrant one. As a result, referring now to FIG. 1c, such an active rectifier provides a forward voltage difference, and associated power loss at a given current, that is much smaller than is achievable with a passive rectifier, while still blocking current flow in quadrants two through four up to the breakdown voltage of the devices used.
FIG. 2 illustrates a circuit model of an ideal active rectifier 200. According to the circuit model, infinite voltage gain feedback amplifier 203, responsive to a voltage difference between first rectifier terminal 201 and second rectifier terminal 202, generates control signal 204, that operates ideal switch 205. Any positive voltage difference between first rectifier terminal 201 and second rectifier terminal 202 results in control signal 204 operating switch 205 to a closed position. As a result, current flows between rectifier terminals 202 and 201 with zero resistance. A negative voltage difference between first rectifier terminal 201 and second rectifier terminal 202 results in control signal 204 operating switch 205 to an open position, thereby blocking all current flow between rectifier terminals 202 and 201.
Contrary to the ideal circuit model illustrated in FIG. 2, in practice, real components have non-zero, forward voltage conduction resistance, as well as finite reverse polarity leakage and breakdown voltage.
Referring now to FIG. 3, a voltage rectifier 303 is illustrated as being incorporated into a “buck” topology DC-DC power converter 300 having DC voltage source 301, MOSFET switch 302, voltage rectifier 303, and an output filter consisting of inductor 305 and capacitor 307. Inductor 305 is normally operated in continuous conduction mode (CCM). Whether voltage rectifier 303 is a passive diode (as shown), or an active rectifier, non-zero minimum load current requirements are imposed.
When switch 302 is ON, voltage source 301 drives VIN to node 304. Current builds up in inductor 305 at a rate determined by the voltage difference between node 304, and node 306, divided by the inductance of inductor 305.
When switch 302 turns OFF, inductor current, IL recirculates through the loop formed by voltage rectifier 303, load 308, in parallel with capacitor 307, and inductor, 305. Current in inductor 305 decreases at a rate determined by the voltage difference between nodes 304 and 306, divided by the inductance of inductor 305. The voltage difference between nodes 304 and 306 equals to VOUT plus the forward voltage drop of voltage rectifier 303. If voltage rectifier 303 is an active rectifier, rather than the passive diode, as illustrated, the forward voltage drop will be substantially lower and the power efficiency of power converter 300 will be improved.
Whether voltage rectifier 303 is passive, with a transfer function illustrated in FIG. 1b, or active, with a transfer function illustrated in FIG. 1c, voltage rectifier 303 permits inductor current in only a single direction: from node 304 to node 306. When current through load 308 is less than the average inductor current, capacitor 307 charges towards VIN, and the output voltage VOUT rises above the time average of the voltage at node 104.
The minimum current required to sustain CCM operation, can be reduced by increasing the inductance of inductor 305. However, increasing the inductance increases the energy storage at any given current level, with the result that the size and cost of the inductor is likewise increased. Increasing the ratio of maximum to minimum current in the inductor also increases the winding resistance, reducing inductor efficiency. Moreover, increasing the inductance increases the characteristic impedance of the output filter, and decreases the output filter bandwidth. Each of these effects increases the size and cost of capacitor 307.
Referring now to FIG. 4, passive voltage rectifier 303 is replaced by a switch 403 operable to perform the current recirculation function of voltage rectifier 303, while also permitting operation in the third quadrant (“reverse current”). For the illustrated circuit, average inductor current can be driven close to zero because inductor current IL reverses for part of each cycle. As a result, supporting a zero average load current condition at node 406 is possible. Referring still to FIG. 4, active switch 403A operates in complementary and mutually exclusive fashion to switch 402. Whereas passive diode 403B only conducts in quadrant one, switch 403A can operate in both quadrants one and three.
A disadvantage of the circuit illustrated in FIG. 4, however, is that switch 403 requires similar overcurrent protection as primary switch 402. It is important in any DC-DC converter applications to protect against excessive current that may damage or destroy circuits. The current in any inductor is the time integral of the voltage applied across it. In a typical buck topology DC-DC converter, the main concern is volt-second product unbalance due to a shorted load. Under these conditions, the volt-second product developed when switch 402 is OFF is very small, and fails to balance the volt-second product developed when switch 402 is ON. Current then rapidly builds in inductor 305. Thus, for the circuit illustrated in FIG. 4, the average on-time of switch, 402, under fault conditions, must be limited to small values.
Referring now to FIG. 5, a known practice is to employ cycle-by-cycle and/or “hiccup” mode current limiting to mitigate the above mentioned problem. A current sensor 509, develops a signal, 510, that is proportional to switch current. A comparator, 513, resets latch 514 when the sensed value exceed threshold voltage 512. Threshold voltage 512 may be selected as varying between zero and a maximum safe value according to an error voltage of a control loop (not shown). Latch 514 turns switch 402 OFF until triggered at the start of the next cycle by synchronization pulse stream 511.
In “hiccup” mode, when a fault level current is detected, switch 402 is switched OFF for a period of time equal to many normal switching cycles so as to limit power dissipation in all components to safe values by limiting average voltage across capacitor 307 and current buildup in inductor 305.
Referring now to FIG. 6, a further known technique is illustrated for protective circuitry useful when the connected load is a dynamic load, for example, when the connected load contains a large energy store such as a capacitor, or is a kinetic load. Here, current sensor 617 is operable to trigger turn-OFF of MOSFET 403A under appropriate circumstances. In addition to latch 514, coupled with switch 402 as described above, a separate latch, 621 is operable to control MOSFET 403A. An OR gate 620 resets latch 621, responsive to the same synchronization pulse stream 511 stream that sets latch 514. Comparator 513 resets latch 514, turning switch 402 OFF. Comparator 513 simultaneously sets latch 621, turning switch 403A on. Typically, additional circuitry (not shown) ensures dead-time between the two switches so as to prevent cross-conduction. Whenever current sense signal, 618, exceeds voltage threshold 622, comparator 619, via OR gate 620, resets latch 621, turning MOSFET 403A OFF. Inductor current diverts through diode 623, back to the input source. Inductor current decays at a rate determined by the difference between VOUT and VIN and the inductance of inductor 305.
Known alternative techniques function similarly, but may utilize a single current sense element that monitors current flow between node 604 and inductor 305. When current flowing from node 604 to inductor 305 reaches a first limited threshold, then the high-side switch, 402, latches OFF. When the current flowing from inductor 305 to node 604, reaches a second limited threshold, the low side switch, 403A, latches OFF.
For the foregoing techniques, employing a synchronous rectifier with overcurrent protection, FIG. 7 depicts the current versus voltage transfer function. A disadvantage of such techniques is that the current sensors 509 and 617, and additional diode 623, represent undesirable additional cost, space, and power consumption.
Referring now to FIG. 8, in a particular application of a DC-DC converter, voltage source 801 is a photovoltaic (PV) substring or module. One or more DC-DC converters within a series string of PV modules connect to an input of an external load, which may be, for example, a shared central power converter, such as a DC to AC inverter 830. Typically, a DC to AC inverter, particularly a one or two-phase output inverter, incorporates a large energy storage capacitor 831, at the input. Capacitor 831 buffers the continuous power delivery from a PV string or array source, from the pulsating power output of the single or two-phase power output inverter. In the absence of diode 825, DC-DC converter third quadrant (reverse) current at 824 would permit discharge of large energy storage capacitor 831. Such discharge can occur rapidly, interfering with operation of inverter 830 and potentially damaging components throughout the attached string. As a result of diode 825, operation of switch 403A in quadrant three can only discharge capacitor 307, and not large energy storage capacitor 831. Thus, diode 825 prevents damaging discharge of capacitor 831. A disadvantage of the above-described technique is that diode 825 is a significant cause of power loss, as well as a contributing undesirably to system cost and size.